Power dissipation products have become prevalent in many testing applications. Generally a power dissipation product functions as an active load drawing a controlled current flow from a power supply in response to a command signal. Thus various significant characteristics of a power supply may be tested by utilizing a power dissipation product which functions as an active load.
In many power dissipation products the controlled current is derived as a function of a voltage command signal. The current generated is proportional to a voltage command signal and may be derived by utilizing a voltage to current converter in a closed loop circuit.
In order to provide an accurate testing system, it is usually preferable to have a power dissipation product free of any oscillations. Consequently the voltage to current converter and the associated circuitry should have less than 360 degrees of voltage phase shift over the entire range of frequencies in which the closed loop circuit has a voltage gain greater than unity.
A preferred implementation of the voltage to current converter uses one or more metal-oxide-semiconductor field-effect transistors, commonly known as MOSFETs. MOSFETs can operate over a wide range of voltages and currents. They also have good low voltage characteristics and very low DC drive power requirements.
FIG. 1 illustrates a typical prior art power dissipation device 10. The power dissipation device includes two load terminals 22 and 24 and operates as an active load. Typically a power supply 12 to be tested is coupled to terminals 22 and 24. The test conductors that connect the terminals of power supply 12 to terminals 22 and 24 exhibit an inductance illustrated as inductances 14 and 16.
The power dissipation device 10 also includes a voltage to current converter 11 which further includes: a MOSFET transistor 36, which receives an error signal from error amplifier 30; feedback resistors 26 and 28, and input terminal 38 which receives a voltage input command signal.
MOSFET transistor 36 operates as a variable current sink for power supply 12. The drain of transistor 36 is coupled to terminal 22. The source of MOSFET 36 is coupled to the current sensing resistor 26 and the feedback resistor 28. Error amplifier 30 is coupled to the gate of MOSFET 36 and receives a voltage input command signal at terminal 38.
Power dissipation device 10 receives a command signal V.sub.1 at terminal 38 and causes a current I.sub.1, the magnitude of which is proportional to V.sub.1, to flow from power supply 12 through the test cable inductance 14, then through the drain of MOSFET 36 to the source of the MOSFET, and then through resistor 26 and the test cable inductance 16 back to a power supply 12. The circuit maintains a constant current I.sub.1 by comparing the voltage across resistor 26, to the current command signal V.sub.1 at terminal 38, and producing via error amplifier 30 an error signal which is provided to the gate of transistor 36. Error amplifier 30 drives transistor 36 further into or out of conduction as necessary in order to keep the voltage across resistor 26 equal to voltage V.sub.1 at terminal 32. Thus a constant current I.sub.1 may be maintained.
However, one disadvantage of MOSFET transistor 36 is the presence of variable parasitic capacitance 34 between the gate and the drain of the transistor. This variable parasitic capacitance may cause oscillation, especially at low voltages where the drain terminal of the MOSFET transistor 36 is operated at voltages near to the voltage at the gate terminal of the MOSFET transistor. The drain to gate capacitance 34 provides undesirable variable magnitude feedback within the converter. The magnitude of the feedback depends mainly on the relationship of drain and gate terminal voltages, and the output impedance of circuitry driving the gate terminal.
The disadvantage caused by the gate to drain variable parasitic capacitance 34, becomes even worse due to the inductances 14, 16 of the test conductors that connect the power supply 12 to the load terminals 22 and 24. The combination of these inductances and the parasitic capacitance increases the probability of oscillation as the value of inductance is increased. This is due in part to the increased voltage swing at the drain terminal of the MOSFET transistor 36 as current is changed.
Oscillation due to the previously mentioned gate-drain capacitance is typically in the order of 10 kHz to 500 kHz, depending on the specific circuit components. Therefore it is difficult to design a wide voltage range electronic load for power supplies. The undesired oscillation prevents a proper testing of the power supply over a desired wide voltage range.
One approach in solving the oscillation problem is the use of a capacitor 18 across the load terminals 22 and 24. The capacitor 18 which has an equivalent series resistance 20, stabilizes the voltage at the MOSFET drain terminals by reducing the voltage swing which is coupled via gate-drain capacitance 34 to the gate terminal. Ideally, capacitance 18 with a small equivalent series resistance 20 is preferred to minimize voltage variations across the drain terminal.
However, one unfortunate result of placing capacitor 18 with small equivalent series resistance 20 across the load terminals 22 and 24 is that the inductances 14 and 16 and capacitance 18 form a resonance LC circuit with a high Q factor. This becomes a serious problem in some types of load operation, for example, when the load is operating in the voltage mode. During operation in the voltage mode, the electronic load controls the terminal voltage of the power supply by continually varying the load current by means of the voltage to current converter which in turn is controlled by the output of the voltage error amplifier which is responsive in turn to the power supply voltage. Consequently, any load current variation generates a ringing in the effective LC resonant circuit which may turn into oscillation due to characteristics of the voltage mode closed loop circuit described above.
Despite the desire for a capacitance with low equivalent series resistance, capacitance 18 has been typically selected with a considerably high equivalent series resistance 20 to provide a damping effect on the ringing caused by the resonant circuit formed by capacitance 18 and inductances 14 and 16. Although the high equivalent resistance would considerably alleviate ringing, it would also cause a wide voltage swing at terminal 22 which is connected to the drain of transistor 36.
The ringing caused by capacitance 18 and inductances 14 and 16 may cause undesirable current overshoots when load current is changed rapidly, for example, during the measurement of the transient response of the power supply. A further disadvantage resulting from the resonance circuit 14, 18 and 16 is that the output of the power supply would experience additional ripple currents and hence actual noise measurements of the power supply could not be accurately attained.
Hence there is a need for a current dissipation circuit with stable voltage characteristic and minimum voltage swing and oscillation across its load terminals.